Adaptive radio frequency (rf) filter

ABSTRACT

The invention relates to an adaptive Radio Frequency (RF) filter ( 11 ), which is particlarly useful as an RF filter in Wireless Local Area Networks (WLAN&#39;s). As greater demands are placed on RF systems, for example in WLAN&#39;s in order to increase channel capacity by utilizing available bandwidth, corresponding demands are placed upon performance and tolerance of components used in FR circuits. An adaptive Radio Frequency (RF) filter for filtering first and second RF signals from an OFDM encoded carrier signal is provided, the adaptive RF filter comprises: a low-pass filter ( 102 ) configured to filter first and second RF signals, one from another, so as to provide a first RF output signal; the adaptive RF filter being tunable in response to one or more input signals, the at least one input signal being derived from a comparator ( 37,100 ), which compares the first RF output signal with a desired value for said first RF output signal, and provides a connection factor for varying a characteristic of the filter ( 102 ). An advantage of the invention is that it facilitates filter of two OFDM encoded RF signals, the first typically at 8.1 MHz and the second (unwanted) at 11.9 MHz, from a base-band signal, with a noise floor level of —55 dB or better. Another advantage is that the filter is able to self calibrate and is able to take into account fluctuations which may affect performance, for example thermal drift, and automatically trim its characteristics so as to compensate for these fluctuations.

FIELD OF THE INVENTION

[0001] The present invention relates to an adaptive Radio Frequency (RF)filter, and more particularly, but not exclusively, to an adaptive RFfilter for use in Wireless Local Area Network (WLAN) transmitters,receivers and transceivers.

BACKGROUND OF THE INVENTION

[0002] RF filters are employed in RF circuits and devices to filterparticular bands of frequencies and/or to remove unwanted or spuriousnoise. As greater demands are placed on RF systems, for example inWLAN's, in order to increase channel capacity by utilizing availablebandwidth, corresponding demands are placed upon performance andtolerance of components used in the RF circuits.

[0003] HIPERLAN 2 is an example of a standard that has been developed inorder to increase channel capacity. HIPERLAN 2 is a Wireless Local AreaNetwork (WLAN) protocol, based on, and incorporating an OrthogonalFrequency Division Multiplexing (OFDM) scheme.

[0004] An advantage of using an OFDM scheme for transmitting RF signalsis that it is possible to transmit more data for a given channelbandwidth than was previously possible. The following modulation formatsmay be incorporated in the HIPERLAN 2 scheme: Binary Phase Shift Keying(BPSK); Quadrature Phase Shift Keying (QPSK); Quadrature AmplitudeModulation (16 QAM) and 64 QAM. The flexibility of use of theaforementioned modulation techniques means that data rates of between 6to 54 Mbit/s can be transmitted.

[0005] A particular advantage of the HIPERLAN 2 scheme, is that systemsincorporating it may be used in offices, shops, airports or in similarenvironments, which were previously prone to multipath dispersion. Thisis because RF transmissions which use HIPERLAN 2 are reflectionresistant.

[0006] Typically HIPERLAN 2 operates at 5.5 GHz with multiple channels;each channel has 52 active sub-carriers within a 20 MHz bandwidth. EightHIPERLAN 2 channels are shown in diagrammatical form in FIG. 3. OFDMmodulation schemes require two component signals to be in phasequadrature. The two component signals are referred to as the I and Qsignals. During operation, and following down conversion to a base-bandsignal, the base-band signal of interest has two 10 MHz side bands, atwhich sub-carrier frequencies, the I and Q component signals lie.

[0007] However, because only a relatively small frequency band separatesadjacent sub-carrier channels, even small changes in filtercharacteristics can cause variations in overall filter behaviour, withthe result that small variations in filter characteristics (for exampleas a result of thermal drift) give rise to cross-talk interferencebetween adjacent sub-carriers. Typically a change in overall absolutetolerance of —16% of the passive components' value prevents the filterfrom achieving the required of —27 dB stop band performance. FIG. 3shows how sub-carriers are separated from a channel. FIG. 4 illustratesdiagrammatically how close together the sub-carrier channels are one toanother.

[0008] It would therefore be advantageous to be able to adjust theroll-off characteristics of two analog low-pass filters so as to meetthe performance requirements specified in HIPERLAN 2.

SUMMARY OF THE INVENTION

[0009] According to a first aspect of the present invention there isprovided an adaptive Radio Frequency (RF) filter arrangement forfiltering first and second RF signals from an OFDM encoded carriersignal, the adaptive RF filter arrangement comprising: a low-pass filtermeans configured to filter first and second RF signals, one fromanother, so as to provide a first RF output signal; the adaptive RFfilter arrangement being tunable in response to at least one inputsignal, the at least one input signal being derived from a comparator,which compares the first RF output signal with a desired value for saidfirst RF output signal and provides a correction factor for varying acharacteristic of the filter arrangement.

[0010] First and second RF signals are preferably adjacent sub-carrierswithin a channel. Prior to filtering the sub-carrier are preferablydown-converted to baseband signals. The down conversion of signals hasthe advantage that lower sampling rates may be used. Filtering ofbase-band signals may be achieved by attenuating channel stop bands by27 dB at a frequency of 11.875 MHz.

[0011] The RF filter arrangement is preferably tuned by varying one ormore capacitive elements. The capacitive elements may be arranged in aparallel network. However, alternative variables may be modified,including without limitation: the resistance of one or more componentsof the filter or the transconductance (gm) of one or more transistor(s)in an operational amplifier used in the filter.

[0012] HIPERLAN 2 (and its US counterpart standard IEEE802.11A) is arelatively new standard. There is no arrangement that performs amatching of two filters to the Wireless LAN requirements using signalsgenerated by an OFDM modem. In particular, no techniques exist forcarrying out timing using a hybrid analog and digital system. Inparicular, no techniques exist for carrying out any timing using ahybrid analog and digital system.

[0013] Preferably a plurality of capacitative elements are connected ina parallel configuration and at least two of said capacitive elementsare arranged to be switched simultaneously so as to maintain asubstantially constant quality factor (Q factor) to the desiredfrequency.

[0014] In one embodiment, the adaptive Radio Frequency (RF) filterarrangment is adapted to vary its frequency response. in accordance witha command signal which is provided by a digital controller. Means areprovided for filtering a first base-band signal from a second base-bandsignal, both signals being OFDM encoded signals. A band-pass filter hasfirst and second outputs, the outputs providing separate signals torespective first and second mixers: A feedback signal is used to derivea signal offset value, whereby, in use, the signal offset value is usedto modify the filter thereby tuning the first signal from the secondsignal.

[0015] Preferably any ripple which may be present in the band-passfilter (0→8.125 MHz) is limited to a maximum value of 1 dB.

[0016] In order to allow the re-construction of I and Q components,first and second filters need to exhibit good matching. That is theyneed to be within ±0.2% of one another.

[0017] Preferably the adaptive (RF) filter arrangement includes: two lowpass filters for 1 and Q signal components, variable capacitors adaptedto modify said low pass filters, means for modifying an OFDM modem togenerate and measure response to a training signal; means for generatinga calibration signal under control of a controller; and a signal path sothat the calibration signal can be applied to the or each filter.

[0018] Advantageously an OFDM digital modem of a wireless LAN (WLAN)system measures a single filter response to a signal, the signalcomprises a plurality of sinusoidal signals, and a controller modifiesor hold the frequency response of the other filter(s); the assumptionbeing made that the other filter(s) have identical characteristics.

[0019] Calibration is advantageously performed during an initialpower-up phase of a WLAN system incorporating the present invention.Alternatively calibration may be performed when the filter isoperational, for example when it is networked with other devices.

[0020] Preferably a reference input signal, which may be derived from aROM look-up table, provides a sample of at least one test signalwaveform.

[0021] The filter response is performed on a single filter output andidentical control is applied to each filter simultaneously. The inputvoltage dynamic range is typically 2 volts peak-to-peak.

[0022] Two frequencies are advantageously used to calibrate the filter.The signals are manipulated by Fast Fourier Transforms (FFT), as theseare readily available. It will be appreciated that more than twofrequencies may be used.

[0023] Advantageously feedback signals are used to calibrate the filter.Stepwise changes are may be achieved using digital feedback signals. Anovershoot capability may be incorporated into a control algorithm so asto permit application of a reverse stepwise increment (decrement), inorder to optimise tuning.

[0024] Calibration includes the function of recalibration. Recalibrationmay be required during operation as a result of, for example, thermaldrift.

[0025] In a particularly preferred embodiment two filters are configuredso that they are multiplexed in transmit or receive modes, therebyexploiting a feature of HIPERLAN 2 scheme, namely that it operates inhalf-duplex mode. That is, information is transmitted in one directionat a time. However, this applies to other standard implementations wheretransmit and receive path filters are independent.

[0026] A preferred embodiment of the Invention will now be described, byway of exemplary example only, with reference to the Figures in which:

BRIEF DESCRIPTION OF FIGURES

[0027] FIGS. 1 to 4 assist in understanding the Invention

[0028]FIG. 1 is a functional block diagram of a known OFDM transceiversystem;

[0029]FIG. 2 is a detailed block diagram of the transceiver of FIG. 1,showing adaptive low pass filters.

[0030]FIG. 3 shows a series of graphs which illustrate diagrammaticallyvarious stages of tuning a single sub-carrier channel from several OFDM(HIPERLAN 2) channels;

[0031]FIG. 4 is a graph of frequency against attenuation and depicts,diagrammatically, two adjacent sub-carrier channels;

[0032]FIG. 5 is a block diagram of adaptive filters in accordance withthe invention;

[0033]FIGS. 6 and 7 are circuit diagrams showing Sallen and Keyconfiguration used in the adaptive filters of FIG. 5;

[0034]FIG. 8 is a circuit diagram of a follower amplifier;

[0035]FIG. 9 is a circuit diagram of a follower amplifier used incascading the filters shown in FIG. 5;

[0036]FIG. 10 is a block diagram of an adaptive filter;

[0037]FIG. 11 is a circuit diagram of the filter in FIG. 10 and showsinput channels F0-F3, from up/down counter, connected to four variablecapacitive networks;

[0038]FIGS. 12 and 13 are graphs showing respectively, theoretical andactual response characteristics of the filters of FIG. 5,

[0039]FIG. 14 illustrates the principle of tuning of filtercharacteristics;

[0040]FIG. 15 is a flow diagram showing key steps in the decision andimplementation of the tuning stages shown in FIG. 14;

[0041]FIG. 16 illustrates diagrammatically the principle of calibrationof filters to compensate for drift;

[0042]FIG. 17 is a circuit diagram of a transceiver including filters inaccordance with the invention.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT OF THE INVENTION

[0043] Referring generally to the Figures, and particularly FIG. 1,there is shown diagrammatically a functional block diagram of an OFDMsystem 10. The system includes an RF transceiver 11 and a base band OFDMprocessor and modem 12.

[0044] Communication between the transceiver 11, processor and modem 12is by way of Digital-to-Analog Converters (DAC) 14, andAnalog-to-Digital Converters (ADC) 16.

[0045] Respective I and Q signals each have separate channels 141 and14Q in the DAC's 14 and 161 and 16Q in the ADC's. FIG. 2 shows in moredetail a block diagram of transceiver 11. Transceiver 11 includes fourfilters 15, 16, 17 and 18. Filters 15 and 16 are band pass filters.Filters 17 and 18 are low pass filters. Each filter 15 to 18 is a 7thorder Tchebychev filter and comprises a cascaded configuration of threesecond order, and one first order, low pass filter sections.

[0046] The quality factors (Q factors) of the second order filters are:5.45, 1.63 and 0.778.

[0047] Detailed diagrams of the filters are shown in FIG. 5. To achievedynamic handling, and to obtain a good signal to noise ratio (SNR), thecascaded configuration of the filters is: 1st order−2nd order(Q=1.63)−2nd order (Q=0.778)−2nd order (Q=5.45).

[0048] As the quality factors are not very high, the so-called Sallenand Key structure is used as shown in FIGS. 6 and 7. Use of thesecomponents helps to keep down the costs of the device. The Sallen andKey structure has other advantages when applied to the presentinvention. For example at low frequencies filter matching depends on thepassive components ratio matching, which is generally very good; and onthe gain of unit gain amplifiers (or follower amplifiers) 19-22.Follower amplifiers 19,21 ensure a gain parameter of around 0.996 in thepass band. Another advantage is the large input signal dynamic that thefollower amplifier 19,21 are able to support. The flexibility of thecascaded configuration therefore allows balancing of signals at theoutput of each 2nd order section.

[0049] To overcome this problem, an up-down counter 100 (See FIG. 10) isused to vary the capacitance of the filters. Based on the outputresponse of an end pass band (frequency=8.125 MHz) sinusoid signals areinjected at the input, the Fast Fourier Transformation (FFT) spectrumanalysis in the OFDM digital modem part governs the up/down controlsignal to adjust the capacitors 23-26 (see FIG. 7).

[0050] The following section describes each component of the filter.

[0051] First, the Sallen and Key structure shown in FIG. 7 requirespassive component ratios (R1/R2 and C1/C2) to be proportional to thesquare of the quality factor Q. By including a unity gain amplifierbetween the two RC circuits, (thereby effectively ‘splitting’ them), theQ factor is proportional to the ratio of the passive components. Thecircuit is shown in FIG. 8 and depicts a symmetrical filter with groundas common mode node. The sensitivity of Q to the gain (K) of a followeramplifier of such a structure is proportional to the square of Q.Typically if Q=5.45 and K=0.9, then ΔQ/Q=(5.45{circumflex over( )}2)*(1−0.9)=3 which is approximately 9.5 dB.

[0052]FIG. 8 shows follower amplifier. Usually, transistor Q1 is usedwithout Q2 and Q3 and produces an error, from the input signal,inversely proportional to the product of the transistor transconductance(gm), and the impedance seen at the emitter of the transistor.Transistors Q2 and Q3, improve the gm 10 times, reducing the error 10orders of magnitude. The gain (K) obtained is 0.996→0.998.

[0053] Q1 of the follower amplifier shifts DC input voltage one VBEdown, causing a loss of 1V at minus 40° C. and reduces the dynamic ofthe filter. NMOS transistor M is used to prevent saturation of Q1. Therelevance of this is explained below.

[0054] To allow the cascading of four sections, a DC voltage shift isnecessary and is achieved by the follower of FIG. 9. Two supplementaryPMOS transistors M1 and M2 are required. There are two outputs: thefirst one is taken at the Q1 emitter which is used to drive feedbackcapacitor C1, the second is connected to the source of transistor M2 andis used to drive the section of the filter. This realizes a DC voltageshift up. PMOS transistors M1 and M2 vary positively with temperature,whereas NPN bipolar transistors vary negatively. All current sources arearranged in such a way that their temperature coefficients compensatethe VBE of the first follower amplifier. Gates of M1 and M2 areconnected together and connected to output of the second R2C2 networkshown in FIG. 7.

[0055] As mentioned above, C1 and C2 are used to tune the cut-offfrequency of filters. The quality factor Q is function of the ratioC1/C2. In order to keep the ratio constant, C1 and C2 must varysimultaneously with the same adjustment for the same frequency shift.The way this is achieved is described below.

[0056] Four bits are provided by digital counter 100 (See FIGS. 10 and11). The least significant bit (LSB) is equivalent to a change incapacitive value of 2%. A diagrammatical implementation of this is shownin FIG. 10. By varying the frequency in this manner, four steps areallowed between one end of a pass-band (at 8.125 MHz) and the beginningof the stop-band (at 11.875 MHz) as depicted in FIG. 14. The controlrange is therefore ±16%.

[0057]FIG. 12 shows examples of the frequency response characteristicsof the filter with 8.125 MHz cut-off frequency. Simulated theoretical(ideal R, C and Unit Gain VCVS) and real responses of each section andof the adaptive filter are shown in FIG. 13. Thus the prediction thatthe tuning can satisfy the specifications is confirmed.

[0058] The transceiver will now be described in operational mode. InFIG. 17, multiplexers 23-26 are switched to calibrate filters 17 and 18.A basic assumption is made here: the characteristics of the filters 17,18 are matched. This means that measurement made on one filter isapplicable to the other, hence any correction(s) made to one filteris/are made on the other filter. This removes the requirement to measurethe characteristics of the other filter(s).

[0059] A calibration input reference signal is issued to the circuitunder calibration. The waveform of this signal consists of a sum of twosinusoidal signals (AL and AH) of known amplitude but differentfrequencies where the filter response to the first signal (ArL) shouldhave no attenuation since (AL) has the frequency of the last active subcarrier carrying information in the OFDM symbol. The signal therefore isnot affected by any roll off of the filter, once calibratedappropriately. The second received signal (ArH) is in the desiredroll-off region of the filter where attenuation is high. The rate ofchange of attenuation with frequency is also important. Signal AH ischosen to be the frequency of the sub carrier with the highest frequency(K=32) in the OFDM symbol previously presented to the filter. In normaldata transfer, in the case of HIPERLAN 2, this is one of the nullcarriers carrying no data information. It corresponds practically to themid position between active sub carriers in between two adjacent bands.

[0060] A reference signal may be generated using a ROM look-up table 27whose contents describe the waveform of the desired time domain signalwhen the contents are read out at the suitable speed. This ensuresdesired precision of the generated signal is in the same order as theprecision of the master clock of the system i.e.: the quartz oscillator(not shown).

[0061] The reference signal is converted to analog via the DAC. Duringthe calibration mode it is then applied to the filter under measurementvia the multiplexers 23-26 to training signal paths 28,29, as shown inFIG. 17, whilst contemporaneously disconnecting signals from a radiotransmit or receive path.

[0062] Other signals are: preset signal, clock signal, an UP/DOWNcontrol signal and an OUTPUT signal from each filter. The filters arecapable of changing their cut-off frequencies as described above bymeans of digital controller 30. Refering to FIGS. 14 and 15;

[0063] Initially, counter 100 is preset (step 31) to maximum value and areference INPUT signal is applied to filter 102 (FIG. 10). The responseof the filter (step 33) is analysed by Fast Fourier Transformer (FFT) 3after digitising the analogue signal (step 33) suitably by ADC ofsufficient dynamic range. Amplitudes of received sinusoidal signals(ArL0 & ArH0), corresponding to the calibration waveform (AL & AH) atthe input of the filter, are stored in memory and serve as “reference”amplitudes for subsequent measurements. Counter 100 is decremented byONE (step 34). This changes the frequency cut-off characteristics offilter 102 and the response of the filter to the same input referencesignal (AL+AH). A new measurement is made after applying an inputreference signal.

[0064] If the magnitude of (ArL2) is within 1 Least Significant Bit(LSB) of the original (ArL0) amplitude of the “reference” signal, a downcount is generated. The process is then repeated until the difference of(ArLn) and the initial (ArL0) is greater than 1 LSB.

[0065] When this situation occurs, the digital part counts UP TWO steps(step 35), measures (step 36) the response of the filter 102 at this newvalue and memorises the values for (ArLn+2) and (ArHn+2).

[0066] Initial calibration may take multiple steps until desired filterroll-off characteristics are obtained. Typically starting just afterreceipt of carrier (AL). Since the FFT (37) has a finite calculationspeed, this may take time that is not in line with such a procedure tobe done in line during an established WLAN session when a link is inplace.

[0067] However, for many reasons, particularly due to temperature drift,an incremental calibration may be required. This can be achieved whenthe WLAN system is in operation, however protocol is such that, timeslots are available when the transceiver is neither transmitting norreceiving. Two OFDM symbol time frames are sufficient to make suchmeasurement and decision. (i.e.: 8 microseconds). Hence incrementalcalibration may be achieved without disturbing data flow.

[0068] In the case of HIPERLAN 2, the most suitable instance is theRandom Access Channel (RCH) time period that is a reserved time in each2 millisecond time frame. Typically the minimum RCH time slot is 9microseconds. Since the modifications of the characteristics of thefilter are incremental, the calibration process in such a case can bemade incremental in order not to exceed the available time slot in RCH.

[0069] At first, the response to the reference signal (AL+AH) ismeasured after a possible drift. In this calibration,(see FIG. 16) onlythe high frequency component of the reference signal is exploited(AriH). If this is greater than δ LSB's ABOVE (ArHn+2), which is thestored value after the initial calibration, then the counter isDECREMENTED by ONE. This is due to a drift that made the filter cut-offdrift towards higher frequencies. If (AriH) is less than δ LSB's BELOW(ArHn+2), then the up/down counter controlling the response of thefilter is INCREMENTED by ONE.

[0070] This is due to a filter that has drifted the cut-off frequency tolower frequencies. Ideally δ is chosen such that the amplitudedifference between two counter steps for the same input frequency can bedescribed in 2δ LSB's (least significant bits) of the output of the ADC.2δ is the smallest change in attenuation for ArH as one changes thesteps to adjust the filter roll-off.

[0071] A digital controller (30) implements control as to how manymeasurement/correction loops are required. In so doing, the digitalcontroller determines the amount of time to be spent in calibration,compared with the amount of time the system has to undertake thecalibration.

[0072] Hence if insufficient time is available in one WLAN time frame,then drift adjustment can be done spread over multiple time frames, onestep at a time. Control and implementation decisions may be carried outunder software control from a link controller (not shown).

1. An adaptive filter arrangement (11) for filtering first and secondsub-carriers from an OFDM encoded carrier signal, the adaptive RF filtercomprising: a low-pass filter configured means (17,18) to filter firstand second RP sub-carriers, one from another, so as to provide a firstRoutput signal, and variable capacitors (C1,C2) adapted to modify saidlow pass filter configured means (17,18); the adaptive filterarrangement comprising: an OFDM encoder (29) for generating acalibration signal under control of a controller (30); a signal path sothat the calibration signal can be applied to the or each low-passfilter configured means (17,18); Fast Fourier Transform means (37) formeasuring a response of the first output signal to the calibrationsignal; and wherein the adaptive filter arrangement is tunable inresponse to one or more input signals, derived from a comparator (30),operable to compares the first response of the first output signal tothe calibration signal with a desired value, and in response to providesa correction factor for varying a characteristic of the filter.
 2. Anadaptive filter arrangement (11) according to claim 1 wherein thecorrection factor varies at least one capacitive element.
 3. An adaptivefilter arrangement (11) according to claim 1 and including two low-passfilters (17,18) for filtering first (I) and second (Q) components from abase-band RF signal.
 4. (Cancelled)
 5. A method of calibrating anadaptive filter arrangement (11) including the steps of; pre-setting(31) an up/down counter generating (32) a reference signal in an OFDMencoder, measuring and storing (33) a filter response to the referencesignal, the measuring including a Fast Fourier Transform decrementing orincrementing (34) the up/down counter, comparing the measured filterresponse with a desired value, and varying a capacitance value dependingupon the comparison, thereby varying the filter response
 6. A methodaccording to claim 5 in which the reference signal is applied to afilter via a training path whilst decoupling a transmit and receivepath.